AN1327 - Very Wide Input Voltage Range, Off-Line Flyback .
AN1327/DVery Wide Input VoltageRange, Off-Line FlybackSwitching Power Supplyhttp://onsemi.comAPPLICATION NOTEAdded to this are those industrial companies which may notonly have their products reside on residential power systemsbut also have the varied international industrial powersystems. This means that a single product family might haveto operate from an input voltage of 90 to 600 VAC, wellbeyond the residential limits of 90 to 270 VAC.This paper reviews one method of enabling adiscontinuous mode flyback converter to operate beyondits traditional range of input voltage of 3:1 to a range of morethan 6.6:1 without affecting the reliability of its operation.This is done by changing its mode of operation and the useof recently available power MOSFETs with breakdownvoltage ratings of 1,200 V.One of the many problems besetting the power supplydesigner today is being able to design a switching powersupply that is able to operate in all the power systems withintheir international marketplaces. Forward mode switchingpower supplies typically operate over a single powersystem’s range of voltage, that is, 90 to 130 VAC or200 to 270 VAC. Boost mode converters can just make therange of 90 to 270 VAC. Any higher input voltages wouldthen require a different design.This leads companies to create products targeted atspecific marketplaces, which can be costly, or to have theircustomers arrange jumpers to accommodate their powersystem which can be annoying or lead to costly errors.Figure 1. The Wide Input Range Flyback Power Supply Demonstration BoardThis document may contain references to devices which areno longer offered. Please contact your ON Semiconductorrepresentative for information on possible replacement devices. Semiconductor Components Industries, LLC, 2012December, 2012 Rev. 21Publication Order Number:AN1327/D
AN1327/DA Summary of the Operation of Fixed FrequencyFlyback ConvertersThe most common topology for those applications lessthan 150 W has been the fixed frequency, current modecontrolled, flyback converter. Its block diagram can be seenin Figure 2. CKPEAK CURRENTCOMPARATORS RORAMPLEADING EDGESPIKESUPPRESSORCURRENTSENSEELEMENTFEEDBACK VOLTAGEINPUT RETURNFigure 2. Block Diagram of a Fixed Frequency Current Mode, Flyback ConverterHere a fixed frequency oscillator initiates a power switchconduction period which is terminated by either the currentwithin the power switch reaching a predetermined limit asset by the error amplifier or the oscillator terminating theperiod and initiating the next power switch conductionperiod.A representative flyback converter can be seen inFigure 3.Tipk DVinVin · TonLpri(eq. 1)The slope of the current ramp is Vin/Lpri.The flyback topology, as with all boost mode converters,operate under the principle of storing energy within the corematerial of the transformer. The energy stored during eachconduction period is given by:Vout The power switch essentially places the primaryinductance of the flyback transformer across the inputvoltage source when it is turned on. The secondary isdisconnected because the output rectifier (D) is reversebiased. The primary winding’s current takes the form of alinear ramp starting from zero amps and whose peak valueis given by:QCONTROLEsto Ipri2Lpri · i pk2(eq. 2)To meet the short term steady state power demands ofthe load(s), the following relationship must be met:Figure 3. A Simplified Schematic of aFlyback Converterhttp://onsemi.com2
AN1327/DPout v2fop · Lpri · i pk2The shortcoming arises in the output drivers of the typicalcurrent mode control IC and the power switch. Typically apower MOSFET is used as the power switch in most modernflyback power supplies. At high input voltages, the on timeof the power switch becomes so short (300 600 nS) thatthe output driver cannot source enough instantaneouscurrent to drive the MOSFET into a saturated conditionbefore turning it back off. The effect is the power switchoperates in the linear conduction mode during these short“on’’ pulses. This causes a drastic drop in power switchoperating efficiency and jeopardizes the power supply’sreliability.(eq. 3)In reality, for any one output power, the current modecontroller strives to maintain a constant value for Ipk over theentire range of input voltages as visualized in Figure 4.PRIMARY CURRENTHI INPUT VOLTAGELOW INPUTVOLTAGEThe New Method of ControlBy a very simple modification to the traditional fixedfrequency current mode controlled flyback converterdesign, one can greatly extend its operational input voltagerange. The modifications make the control method one ofvariable on time, and variable frequency. Figure 5illustrates the newly added and redefined functional blocksof this new method of control.IpkTIMEFigure 4. Peak Currents at Differing Input Voltages DRIVERS IME DELAYCURRENTSENSEELEMENTFEEDBACK VOLTAGEINPUT RETURNFigure 5. Block Diagram of the Wide Input Range, Flyback Converterentire output voltage swing. The other new block is really aredefinition of an old familiar function the leading edgespike filter from the current sensing element. Here, theformerly annoying parasitic of time lag serves an importantfunction within the control algorithm. It now delays theactual current ramp prior to being sensed by the control IC.It allows the actual peak current to increase with increasinginput voltages while the controller sees a lowering peakcurrent needed by this control strategy. This will bediscussed later.A VCO (voltage controlled oscillator) is created byremoving the timing capacitor’s charging circuit from afixed voltage or current source and placing it under thecontrol of the error voltage. In the design example shownlater, it means simply removing the timing resistor from thevoltage reference and wiring it to a variable voltage createdby the output of the error amplifier. A voltage translator isplaced between the output of the error amplifier and thecontrol input to the VCO. It consists of a simple biased 3.3 Vzener diode so that the error amplifier may make use of itshttp://onsemi.com3
AN1327/Dfixed frequency flyback converter. This will allow us todetermine the appropriate value for the primary inductance.In the sample design, the frequency of operation at thehighest input voltage will drop to one half from that at thelowest input voltage. Equation 4 then dictates:The ultimate goal of the new control methodology is toforce the on time of the power switch to be greater than thisminimum effective on time over the power supply’s entireline/load operating range. Its operation can be bestunderstood by examining equation 3. The erroramplifier/VCO section of the circuit lowers the operatingfrequency as the input voltage is increased. This requires theenergy stored per conduction period to increase to meet theshort term power requirement of the output. This is done byextending the on time of the power switch. If the keycomponent parameters such as maximum operating fluxdensity (Bmax) of the transformer, the avalanche ratings ofthe diodes and power switch, and the current ratings of theoutput rectifiers are adequate, then no degradation in thereliable operation of the supply is experienced.Its operation can be better defined by rearrangingequation 3 and neglecting any power loss due to theinefficiency of the supply one gets:ipk Ǹ2 PoutLpri · f(f(Ve))ipk(hi) [ Ǹ2 ipk(lo)If the desired maximum operating flux density (Bmax) isone half the core material’s saturation flux density at 100 Cand at the high input line, then the operating flux density atthe low input voltage should be:Bmax(lo) [Ǹ2 Pout · Lprif(f(Ve))Bsat(min)2 Ǹ2(eq. 7)For most common ferrite materials such as 3C8, N27, orF, the operating flux density at low line will be atapproximately 1,300 gauss. The Bmax at the high inputvoltage will be no more than one half the saturation fluxdensity at 100 C.The inductance can now be calculated by using equation 1at the low input voltage, and an air gap calculated using anyone of the common methods.It is important to determine the secondary inductance suchthat the core’s energy can be emptied as close to 50 percentduty cycle (1/fop(hi)) as possible. This will minimize theRMS currents to their lowest possible point over the entireoperating range. The output peak current at any operatingpoint is described as:(eq. 4)where f(f(Ve)) is the controlled frequency of the power supply.As one can see, the peak current is inversely proportionalto the square root of the frequency of operation, since all theother terms are fixed in the short term operation and by thecircuit design.By substituting equation 1 into equation 4 one further gets:ton 1 ·Vin(eq. 6)ipk(out) [ 2 · Iout(av) · Tdisch · fop(eq. 5)(eq. 8)This would describe both the peak currents flowingthrough the output rectifiers and the peak ripple currentsflowing into and out of the output filter capacitors. Withinthe sample design with one ampere rated outputs, at low linethe peak to peak rectifier currents would be four times theaverage output current. At the high line, the peak to peakcurrents would be eight times the average output current forthe rated output current.There are more unknowns than there are independentequations, but at the low input line voltage and at the ratedoutput load, one can solve equation 5. The input voltage isknown to be 125 VDC (90 VAC), the frequency will be at itshighest point as designated by the designer, the on time willbe one half of the entire operating period and the peakcurrent will be calculated as it is in a commonhttp://onsemi.com4
AN1327/DThe Wide Range Flyback Converter Demonstration BoardThe wide input range off line flyback converter described has the following maximum and performance ratings.Output Power:17 WattsOutputs: 5.0 Vdc @ 1.0 Amp Max 12 Vdc @ 1.0 Amp MaxInput Voltage Range:90 VRMS 600 VRMSMaximum Input Voltage:675 VRMSJ1D1C2L1R1R2R9D3 H2C1 D4C3 C4C5R5R16D5U1R12C7C8D10R6 T1C11D8D9 C12Q1GC10WIDE INPUT RANGEFLYBACK POWER SUPPLYITC109BR7C131D6R10R8C6C9CAUTIONHIGH VOLTAGESINDUSTRIALTECHNOLOGY CENTERR11R4D2H1GNDR3C16R13R14R15 C17C15D7R20U2Figure 6. Printed Circuit Board Layouthttp://onsemi.com5R17 C14 R18R19J2O/P 12V 5VU3R21GND
AN1327/DTable 1. Parts ListDesignatorQuantityValue/RatingC1, C420.1 mF, 1.0 kVCapacitor, CeramicDescriptionC2, C320.0047 mF, 3.0 kVCapacitor, CeramicC5, C62100 mF, 450 VCapacitor, ElectrolyticC71220 pF, 50 VCapacitor, CeramicC811000 pF, 50 kVCapacitor, CeramicC911000 pF, 1.0 kVCapacitor, CeramicC10110 mF, 25 VCapacitor, TantalumC11, C122100 mF, 20 VCapacitor, TantalumC13, 142100 mF, 10 VCapacitor, TantalumC1511500 pF, 50 VCapacitor, CeramicC1611.3 mF, 50 VCapacitor, MylarC1710.22 mF, 35 VCapacitor, CeramicD1 D441.0 A, 1.0 kVDiode, 1N4007D513.3 V, 500 mWZener Diode, 1N5226BD611.0 A, 1.0 kVDiode, UF, MUR1100ED714.0 A, 300 VDiode, UF, MUR430ED813.0 A, 70 VDiode, Schottky, MBR370Diode, UF, MUR130ED911.0 A, 300 VD101 Diode, SIGNAL, 1N4148J11 ConnectorJ21 ConnectorL11 Coil Craft P/N E3493Q113.0 A, 1.2 kVR1 R44470 kW, 1/2 WResistorR5 R9582 kW, 1/2 WResistorR1011.8 kW, 1/4 WResistorR11127 kW, 1/4 WResistorR12110 W, 1/4 WResistorR1311.0 kW, 1/4 WResistorR1411.2 W, 1/2 WResistorR151680 W, 1/4 WResistorR161100 kW, 1/2 WResistorR1717.5 kW, 1/4 WResistorR1813.57 kW, 1/4 WResistorR19132.4 kW, 1/4 WResistorR201120 W, 1/4 WResistorR2112.49 kW, 1/4 WResistorT11TransformerU11 IC, UC3845BU21 Optoisolator, MOC8102U31 IC, TL431CLPhttp://onsemi.com6MOSFET, MTB3N120ECramer, CSM, 3015 027
AN1327/DL1H1C190 VAC 0.1600 VAC 1 kVD1 D41N4007sC40.11 kVL1 VinC6100 mF450 VH2C30.00473 kVEARTHGNDC20.00473 kV C5100 mF450 V R4470 k1/2 WR3470 k1/2 WR2470 k1/2 WR1470 k1/2 WINPUT GNDFigure 7. AC Input/Filter Circuit SectionT1D9MUR430C11D8100 mFMBR37010 V VinVauxR982 k, 1/2 WR8100 mF20 VR7R6R5D10R16100 k1/2 W10 mF25 V LLR111.8 kD53.3 V641C7220 pFC9C13MUR130MUR1100D6D77UC3845BNR1027 k1 nF3 kV2U21/2MOC810253MTB3N120ER12 10 W C81000 pFC14VauxR141.2 W1/2 WINPUT GNDFigure 8. DC/DC Converter Circuit Section7C12U2MOC8102C172.2 nFQ1http://onsemi.com 12V 5V R131kR15680 W R20120 WC151.5 nFR1932.4 k1.3 mF 7.5 kU3TL431C16R17R212.49 kGND
AN1327/DDesign of the Wide Input Range FlybackConverterNpri 1000Predesign Considerations 1000Output Power:Po (5.0 V)(1.0 A) (12 V)(1.0 A) 17 WattsDC Input Voltages:Vin(low) 1.414 · Vin ac(low) 1.414(90 VAC) 127 VDCVin(hi) 1.414 · Vin(hi) 1.414(600 VAC) 854 VDCMaximum Average Input:Current:Iin av(max) Pout/(eff · Vin(min)) (17 W)/(0.8)(127 VDC) 167 mAN sec N() 5) (eq. 9)(eq. 10)Co [The minimum length of the airgap for the core is then: Io(max)fest(min) · Vripple(max)(1 A)[ 142 mF(70 kHz)(0.1 V)280.4 p Lpri i pk10AC B2max(eq. 14)Sizing the Output Filter CapacitorsSince this is a variable frequency system, all thecalculations for the value of the output filter capacitors willbe done at the lowest frequency since the ripple voltage willbe greatest at this frequency. Since capacitor values aredetermined by the output current, and both the 5.0 V andthe 12 V outputs have the same maximum output currentrating, their capacitance values will be equal.Using equation 6, one gets the maximum operating fluxdensity at the low input voltage of:Ig (5 V ) 0.5 V)(8T) 4 turns(12.9 V)The amount of error between the actual transformeroutput voltages and the required output voltages are: 5 V: 5.0 V 12 V: 10.1 V after the rectifier drop.Add one turn (9 turns) to the 12 V outputs. The resultingoutput voltage, including the rectifier drop is 11.5 volts.The physical winding of the transformer is extremelyimportant (refer to Figure 9). First, there are the creepagerequirements (space between windings over surface) of thesafety agencies. Secondly, with 850 volts across the primarywinding at the high input voltage, the interlayer voltagecould cause arcing between layers of the primary winding.A layer of Mylar tape must be placed between adjacentlayers of the primary. The final transformer construction isgiven below.0.5(127 V) 553 mH(0.82 A)(140 kHz)3500 G 1237 G2 Ǹ2(eq. 13)The auxiliary winding to power the control IC is also 12 V, so it will have the same number of turns.The number of turns needed for the 5.0 volt winding is:Designing the TransformerAfter reviewing the core sizing information provided bythe various core manufacturers, it is decided that an E Ecore of about 1.2 inches (30.5 mm) on a side will adequatelyfit the windings and insulation needed by this application.This corresponds to a Magnetics Inc. part number43007 EC core (or Philips 782E272 (E 30)). The corematerial should be a Magnetics P, F or N material or Philips3C85 or 3F3 material. A Magnetics part numberF 43007 EC will be used.Calculating the primary inductance needed for thisapplication:Bmax(lo) [(Vout ) Vfwd)(1 * ē max) · Npriē max · Vin(min)(12 V ) 0.9 V)(0.5)(74T) 7.5 turns, make 8 turns(0.5)(127 VDC)Nominal Peak Current:Ipk 5.5 · Pout/Vin(min) 5.5(17 W)/127 V 0.74 AmpsThe desired maximum frequency of operation is 140 kHz.ē max Vin(min)Ipkf maxǸ(eq. 12)0.553 mH 74.4 turns, make 74 turns100 mHThe number of turns needed by the 12 V secondary, andassuming ultrafast recovery rectifier is:NOTE: The primary winding’s AWG should be #30 AWG.Lpri ǸLApriL(eq. 15)Make the output capacitors two 100 mF capacitors placedin parallel for each output (C11 and C13, C13 and C14).(eq. 11)0.4 p (553 mH)(0.82 A)2 108 0.046 cm or 18 mils(0.6 cm2)(1300 G)2Designing the Voltage Feedback SectionThe internal error amplifier in the UC3845 (U1) will notbe used. The inverting input pin should be grounded toensure that the output will be always high. The erroramplifier function will be provided by a TL431 (U3) on theAn airgap that produces an AL of 100 mH/1000T is largerthan this, so that is what is used.The number of turns needed to produce the requiredprimary inductance is:http://onsemi.com8
AN1327/Dsecondary being connected to the primary side via anoptoisolator, the MOC8102 (U2).The collector of the MOC8102 optoisolator represents thekey control node for power supply. The value of the voltageat this node sets both the frequency of operation and the peakcurrent flowing through the power switch during each cycle.The collector of the U2 will be connected to thecompensation pin of the UC3845 which will directly set thepeak current. Then a 3.3 volt zener diode (D5) will elevatethis voltage to a higher voltage to set the frequency of thevoltage controlled oscillator. Choosing the maximumcurrent from the output of the MOC8102 optoisolator to be5.0 mA, an external resistor (R11) from the VCC of the IC tothe VCO input is needed. Its value is set when theMOC8102’s output is at saturation and is:R11 (12 V * 3.3 V)ń5 mA 1740 ohms, make 1.8 k(eq. 16)The MOC8102 has a Ctrr of 100 percent. That makes theLED current 6.0 mA (5.0 mA from R10, 1.0 mA from pin 1of U1). A margin of 30 percent should be added forvariations in the gain of the optoisolator. That would makethe LED current 8.0 mA.The value of the current limiting resistor for theoptoisolator LED (R20) is:R20 [5 V * (VU3 ) VLED)]ń8 mA 138 ohms, make 120 k(eq. 17)BOARDER TAPE5 mm 5 V AND 12 V WINDINGS(SELF RATED)3 LAYERSTAPE EA. 12 V WINDING (Vaux)KAPTON INSULATED1 LAYERTAPEPRIMARYBOARDER TAPE4 mmFigure 9. Construction of the Transformerbe at its highest and the duty cycle will be 50 percent. TheVCO control node will be at its highest linear value whichis 7.7 volts.One starts by selecting the size of the timing capacitor (Ct,C7). This is done by referring to UC3845 data sheet,Figure 2 “deadtime vs. frequency.’’ It is desired that thedeadtime be a minimum, since the UC3845 is already 50percent duty cycle limited. At an oscillator frequency of280 kHz, which is divided by 2 for an operating frequencyof 140 kHz, the largest capacitor that yields the leastdeadtime is approximately 220 pF.Using Figure 1 from the UC3845 data sheet “TimingResistor vs. Oscillator Frequency’’ and knowing the VCOcontrol voltage will be 2.2 V higher than the 5.0 Vreference assumed by the chart, one can “scale’’ Figure 1 sothat the same charging current is flowing through the timingresistor (R10) but from the higher voltage source. So bymultiplying the ratio of 7.7 V divided by 5.0 V by the valueof the resultant resistor value from Figure 1, one gets theapproximate final resistor value. Figure 1 results in a valueof 18 k ohms for the timing resistor for a timing capacitor of220 pF. The final value of R10 is then:The lower resistor of the voltage sensing network (R21)is set by assuming a sense current. One milliamp yields 1.0 kohm per volt, which is easy. So:R21 VrefńIsense 2.5 Vń1 mA(eq. 18) 2.5 k make 2.49 k 1%Splitting the output voltage sensing between more thanone output will improve the cross regulation of all theoutputs. The 5.0 V output is usually connected to MCUswhich are voltage sensitive. Usually, the loads connected tothe 12 V output are less susceptible to voltage variations.Select the proportion of sense current to be 70 percent fromthe 5.0 V and 30 percent from the 12 V outputs. The valueof the 5.0 V sense resistor (R18) is:R18 (5 V * 2.5 V) 3.57 k, 1%0.7(1 mA)(eq. 19)The 12 V sense resistor (R19) is:R19 (12.2 V * 2.5 V) 32.3 k0.3(1 mA)(eq. 20)make it 32.4 k, 1%Designing the Voltage Controlled OscillatorOne designs the VCO component values when the powersupply is at the lowest input voltage. Here the frequency willRt http://onsemi.com97.7 V (18 k) 27.7 k make 27 k5.0 V(eq. 21)
AN1327/DThe lowest frequency that the power supply is capableoccurs when the error amplifier is at its lowest output whichis about 0.8 V. The lowest operating frequency would thenbe:flow Rst PD The Current Sense ResistorDetermining the value of the current sense resistor (R14),one uses the peak current determined in the predesignconsiderations and at the minimum input voltage. To keepthe current mode operation linear, the peak currents must bekept less than 1.0 volt in normal operation. So to find thecurrent sensing resistor (R14) value:The Voltage Feedback Loop CompensationThe output that is most heavily sensed is the 5.0 Voutput, so that will be the output that is used as the referenceinput for the feedback loop analysis.The output filter pole at light load (0.1 A) of this output is:(eq. 23)1ffp() 5) 2pRoCo(eq. 27)1 15.9 Hz2p(50 W)(200 mF)The Current Ramp Time Delay CircuitThis circuit is very important to the operation of theoverall circuit. Aside from providing the usual spikeelimination to the current comparator, it also provides a timedelay function from the current sense resistor to the input tothe current comparator. Although a wide range of resistorand capacitor values will work, some minimum time delayis required to avoid instabilities due to too short an on timeat the high range of input voltages.One starts by selecting a value for the capacitor (C8). Thecommon range of values for this function are 470 pF to1000 pF. For this particular application a value of 1,000 pFwas selected for C8. A good estimate of the time delayneeded would be approximately 0.7 mS. This is also theapproximate minimum on time at the high input voltage.The value of the resistor (R15) would then be:(700 nS) 700 W make 680 ohms(1000 pF)(Vin(max))2(854)2 1.66 W (eq. 26)Rst4(110 k)This makes the power dissipated in each resistor 0.41 watts.This is a little high, so place five resistors (R5 R9) eachhaving a value of 82 k ohms and each will dissipate 0.36watts which is below a 25 percent derating point.1.0 VVRsc sc 1.35 W make 1.2 ohms, 1ń2 WIpk0.74 A (eq. 25)Each resistor is then 110 k.The power dissipated is:(0.8 V ) 3.3 V)· 140 kHz 75 kHz (eq. 22)7.7 VTRf dCfVin(min)127 V 423 kIstart(min)0.3 mAThe 5.0 V output filter pole at rated load (1.0 A) is159 Hz.The zero contributed by the ESR of the output filtercapacitors will be approximately 15 kHz.The gain exhibited by the open loop power supply at thehigh input voltage will be:ADC (Vin * Vout)2 N secVin · Ve · Npri(854 V * 5 V)2 4T 48.3(854 V)(1)(70T)(eq. 28)or GDC 33.6 dB. This is the highest DC gain that will beexhibited by the open loop power supply and will reduce to16.5 dB at the low input line. This will reduce the bandwidthof the closed loop power supply by almost a decade whengoing from high input line to low input line. This ismarginally acceptable. By setting the widest allowablebandwidth at the high input line, then one can be assured ofa reasonable bandwidth at the low input line. The maximumrecommended bandwidth is approximately:(eq. 24)The Start Up CircuitA passive start up circuit is used. That is, resistors willbring current from the input line to start up the control IC.It is desired that a “hiccup’’ mode of overcurrent protectionbe implemented, which means that the amount of currentthat flows through the start up circuit must be less than thecurrent needed to run the control IC. The UC3845B usesapproximately 10 mA during normal operation and drawsbetween 0.3 to 0.5 mA in standby. The start up energy willbe stored in the 10 mF filter capacitor.To meet the breakdown rating of the half watt resistors(approximately 250 V), a minimum of four resistors inseries will be needed to accommodate the 854 VDCmaximum input voltage. The total start up resistance willbe:fxo fsw(min)75 kHz 15 kHz55(eq. 29)The gain needed to be contributed by the error amplifierto achieve this bandwidth is calculated at rated load becausethat will yield the widest bandwidth condition which is:ǒffxofp Ǔ * GDC15 kHzǓ * 33.6 dB 5.9 dBGxo 20 Log ǒ159 HzGxo 20 LogThe gain in absolute terms (needed later) is:http://onsemi.com10(eq. 30)
AN1327/DAxo 10ǒG20xoǓǒ Ǔfc fsw · 10 Att40(eq. 31)Axo 10(5.9 dBń20) 1.97where Att is the attenuation needed at the switchingfrequency in negative dB.Now the compensating circuit elements can be calculated.12 · p · Axo · R19 · fxo1 2 · p (1.97)(3.57 k)(15 kHz)ǒ(eq. 32) (1.97)(3.57 k) 7033 ohms, make R20 7.5 kThe compensating zero must be placed at or below thelight load filter pole.12 · p · R20 · fze1 1.3 mF2 · p(7.5 k)(15.9 Hz)L (eq. 33)C (50)(0.707) 598 mHp (18.8 kHz)(eq. 37)11 (2pfc)2L[2p (18.8 kHz)]2(598 mH)Post Design ModificationsIt was found that the control circuitry drew more quiescentcurrent than anticipated. That made the start up voltagehigher than desired. It was also found that the optoisolator’sdark collector leakage current and the timing capacitor’sleakage current were responsible for this behavior. D10 andC17 which isolated these elements behind a low reverseleakage diode during the start up process were added.Toff · Iin * av(max)(5 mS)(0.167 A)[ 42 mF20 Vp · fcCoilcraft offers off the shelf common mode filterchokes (transformers) and the part number closest to thisvalue is E3493. With this filter design a minimum of 40 dBbetween the frequencies of 500 kHz and 10 MHz can beexpected. If later during the EMI testing stage, additionalfiltering is needed, a third order to the filter design will beadded by using a differential mode filter.The Bulk Input Filter CapacitorThe approximate value of capacitance needed is:VrippleRL · z 0.1 mFDesign of the Input Rectifier/Filter CircuitThis circuit provides EMI filtering, rectification and bulkenergy storage for the power supply. It has some severeoperating conditions it must withstand, such as very high ACand DC voltages at the high input voltage range. Highvoltage ratings for the rectifiers and bulk filter capacitors areneeded. Also large creepage distances, the distance an arcmust travel over a surface, must be maintained to meet therequirements of the safety regulatory agencies.1N4007’s will be used as the input rectifiers because oftheir 1000 V reverse voltage ratings and the average inputcurrent of the power supply is less than 1.0 amp.Cin [(eq. 36)A damping factor of 0.707 or greater is good and itprovides a 3.0 dB attenuation at the corner frequency anddoes not produce ringing in the filter reactances. Assumethat the input line impedance is 50 ohms since the regulatoryagencies use a LISN (Line Impedance StabilizationNetwork) which makes the line impedance equal this value.Calculating the values needed in the common modeinductor (L1) and “X’’ capacitors (C1 and C4):R17 Axo · R18C16 Ǔfc (75 kHz) 10 * 24 18.8 kHz40C15 1500 pF(eq. 35)(eq. 34)Performance of the Sample DesignOutput Regulation: 5.0 V 1.2% 12 V 1.5%make this two 100 mF, 450 VDC capacitors in series (C5 andC6).Input Regulation: 5.0 V 0.4% 12 V 0.5%The EMI FilterA second order, common mode filter is used. The lowestfrequency of operation occurs at the low input voltages. Theestimated lowest frequency of operation is 75 kHz. This isimportant to attenuate the switching noise sufficiently topass EMI testing.A good starting point is to assume that 24 dB ofattenuation at 75 kHz is needed. That makes the cornerfrequency of the common mode filter as:Output Ripple: 5.0 V100 mV (@ 90 VAC) 12 V130 mV (@ 90 VAC)http://onsemi.com11
AN1327/D1603.51403.01202.5ON TIME (mS)FREQUENCY (kHz)Figures 10 through 13 graphically represent some of the parameters of the evaluation board that are important toits operation. These parameters were measured at full rated load for all of the 0400500600700800INPUT VOLTAGE (VDC)Figure 10. Frequency vs. Input VoltageFigure 11. On Time vs. Input Voltage0.60.60.50.50.40.30.20.10120200INPUT VOLTAGE (VDC)PEAK CURRENT (A)PEAK CURRENT (A)1.50.52001202.0854ACTUAL PEAK CURRENT0.4MEASURED PEAK 400500600700800854INPUT VOLTAGE (VDC)INPUT VOLTAGE (VDC)Figure 12. Actual Peak Current vsInput VoltageFigure 13. Actual and Measured Peak Currentvs. Input VoltageON Semiconductor andare registered trademarks of Semiconductor Components Industries, LLC (SCILLC). SCILLC owns the rights to a number of patents, trademarks,copyrights, trade secrets, and other intellectual property. A listing of SCILLC’s product/patent coverage may be accessed at www.onsemi.com/site/pdf/Patent Marking.pdf. SCILLCreserves the right to make changes without further notice to any products herein. SCILLC makes no warranty, representation or guarantee regarding the suitability of its products for anyparticular purpose, nor does SCILLC assume any liability arising out of the application or use of any product or circuit, and specifically disclaims any and all liability, including withoutlimitation special, consequential or incidental damages. “Typical” parameters which may be provided in SCILLC data sheets and/or specifications can and do vary in different applicationsand actual performance may vary over time. All operating parameters, including “Typicals” must be validated for each customer application by customer’s technical experts. SCILLCdoes not convey any license under its patent rights nor the rights of others. SCILLC products are not designed, intended, or authorized for use as components in systems intended forsurgical implant into the body, or other applications intended to support or sustain life, or for any other application in which the failure of the SCILLC product could create a situation wherepersonal injury or death may occur. Should Buyer purchase or use SCILLC products for any such unintended or unauthorized application, Buyer shall indemnify and hold SCILLC andits officers, employees, subsidiaries, affiliates, and distributors harmless against all claims, costs, damages, and expenses, and reasonable attorney fees arising out of, directly or indirectly,any claim of personal injury or death associated with such unintended or unauthorized use, even if such claim alleges that SCILLC
A representative flyback converter can be seen in Figure 3. Figure 3. A Simplified Schematic of a Flyback Converter CONTROL Vin T Q Ipri Vout D The power switch essentially places the primary inductance of the flyback transformer across the input voltage source when it is turned
Primary nominal voltage 28 Vdc : Maximum voltage 32.2 Vdc : Minimum voltage 22.0 Vdc Emergency voltage 18.0 Vdc POWER INPUT 'off' voltage 15.0 Vdc Input current at Emergency voltage 1.0 A . 1.4.1.1 Audio Performance Rated Input Level SPEAKER INPUT audio rated input level 1.00 Vrms 10% . SPEAKER MIC input level 250 mVrms 10%
Amplifier Configurations . Voltage Amplifier: Voltage input and Voltage output The controlled source is a Voltage -controlled-Voltage Source A. v Open Circuit Voltage Gain can be found by applying a voltage source with R. s 0, and measuring the open circuit output voltage(no load or R. L infinity) Source Amplifier Load . Amplifier Input .
external drain shield to the appropriate earth ground on one end. INPUT BOARD Input 3 Common Input 3 Input 4 Common Input 4 Input 1 Input 1 Common Input 2 Input 2 Common Relay 1 NO Relay 1 C Relay 1 NC Chassis GND At Other Panel 1 2 3 485-Com 485-Com - Com-Gnd RS 485 Com Bus Cable From Diff. Panel (Remote Mount Only) Power J1 - RS485 .
output voltage to input voltage depending on number of stages is called Cockcroft-Walton Voltage multiplier circuit which is used to develop high voltages in order of several kV. An output voltage, from any stage, can be taken out through tapings. In this work, the input voltage, for Cockcroft-Walton voltage multiplier, has been taken
input and the output voltages. The input voltage must remain typically 2.0 V above the output voltage even during the low point on the input ripple voltage. XX, Input MC78XX Cin* 0.33 F CO** Output These two digits of the type number indicate nominal voltage. Cin is required if regulator is
38 - Check voltage output to fuel pump @ relay set 39 - Check voltage input to relay set from ignition switch 40 - Check voltage input to fuel pump relay of relay set 41 - Check input voltage to control relay of relay set 42 - Check input voltage to control relay of relay set (continued)
is the input voltage plus the transformer secondary voltage, which is equal to the input voltage divided by the transformer turns ratio. By varying the PWM duty cycle from 0 to 100% the average output can be controlled between a minimum of the input voltage, and a maximum of the input voltage plus the boost provided by the transformer secondary .Cited by: 1Publish Year: 2003Autho
The Nutcracker Ballet is derived from the story “The Nutcracker and the King of Mice” which was written E. T. A. Hoffman. The story begins on Christmas Eve in 19th Century Germany. It begins in the Stahlbaum’s house where everyone is preparing for their festive Christmas Eve party. The Stahlbaum’s house is a large and beautiful home, with the grandest Christmas tree imaginable. Mrs .
