Photodiode Amplifier With Transimpedance And Differential .
15th International Conference on DEVELOPMENT AND APPLICATION SYSTEMS, Suceava, Romania, May 21-23, 2020Photodiode Amplifier with Transimpedance andDifferential Stages for Automotive Visible LightApplicationsCătălin Beguni1,2, Alin-Mihai Căilean1,2,, Sebastian-Andrei Avătămăniței1,2 and Mihai Dimian1,212Department of Computers, Electronics and Automation, Stefan cel Mare University of Suceava, 720229 Suceava, RomaniaIntegrated Center for research, development and innovation in Advanced Materials, Nanotechnologies, and Distributed Systemsfor fabrication and control, Stefan cel Mare University of Suceava, Suceava, RomaniaAbstract— To address the most common issues found inautomotive Visible Light Communication applications, a newphotodiode amplifier based on an instrumentation amplifier isproposed. In order to improve the results, buffer stages havebeen replaced with transimpedance amplifiers providing thecircuit with offset cancellation loops that enable the sensing of theDC output offset and with T-networks in the feedback path, thatreduce the effect of parasitic capacitances. Thus, the resultedoutput signal will be ready for further processing in the analogdigital conversion (ADC) stage providing improved results interms of parasitic light resilience and SNR.Keywords—automotive applications, common-mode rejectioncircuit, transimpedance amplifier, visible light communications.I.INTRODUCTIONVisible Light Communications (VLC) are a relatively newtechnology that uses the visible light spectrum (380-780 nm)and the fast switching ability of solid-state lighting devices forlighting and wireless data transmission simultaneously [1], [2].Due to the wide distribution of LED lighting sources whichmakes the VLC technology already half-implemented worldwide, numerous applications have already been found. Thus,the VLC technology is suitable for high data rate indoorapplications [3], [4], high precision indoor localization [5],communication based vehicle safety applications [6], intervehicle distance determination [7], underwater communications[8], medical applications [9], while numerous otherapplications are continuously emerging. Nevertheless, in orderto provide all these services, numerous challenges still need tobe addressed. Many of these challenges are related tocommunication protocols, modulation and coding techniques,multi-user access and resource sharing, whereas some are stillassociated to the physical layer. Thus, the further improvementand development of high performances transimpedance circuitsrepresents a hot research topic, as this stage has the highestimportance in determining the overall performances of VLCsystems [10]-[13]. So, in order to further expand all of theabove mentioned applications, improved transimpedancecircuits have to be developed continuously.In this context, this paper proposes a new design oftransimpendace circuit aimed for automotive applications [6].This work was supported by a grant of the Romanian Ministry ofResearch and Innovation, CCCDI - UEFISCDI, project number PN-III-P33.1-PM-RO-FR-2019-0282 / 21 BM 2019, within PNCDI III.The proposed design aims to provide high resilience to noiseand high gain-bandwidth product (GBP). The paper describesall stages concerning the design and development of theproposed transimpedance circuit, providing this way a usefultool for such designs.II.CONSIDERATIONS ON THE DESIGN REQUIREMENTSA VLC receiver mainly consists of several main blocks: anoptical collecting system with lens, optical filters and aphotodetector, a preamplifier, a filtering stage, an amplifierstage along with clock and data recovery. Additionally, insome cases, biasing and compensation blocks may benecessary. The conversion of light into electric current can bedone with various optoelectronic devices. However,photodiodes are singled out due to low cost, low switchingtime, good frequency response, linearity, low noise and highsensitivity reasons. However, the current generated by aphotodiode is too small to be processed directly and so, it isrequired to convert it into a signal that is easier to process [13].This conversion is done at the preamplifier stage and ideally, itshould be done while limiting the amount of generated noise.As the preamplifier represents the main source of noise in theentire amplifier chain, its design will determine the sensitivityof the receiver.There are three main possible implementations: with a lowimpedance amplifier, with a high impedance amplifier and witha transimpedance amplifier (TIA), each with its ownadvantages and disadvantages. The low-impedance amplifierhas the widest input range, so a very good bandwidth can beachieved easily. However, this solution has as disadvantage alower Signal-to-Noise Ratio (SNR), due to a small-valueresistance at the input. In order to obtain a high SNR, a highimpedance amplifier can be used instead. This solutioninvolves the usage of a high-value resistance, leading to anearly saturation of the amplifier due to higher input levels. Thetransimpedance amplifier is a compromise between the firsttwo solutions, as it is able to provide medium noise, highsensitivity and high bandwidth. These characteristics areachieved due to the fact that it typically has wider input currentranges than high-impedance amplifiers and better sensitivitythan low-impedance amplifiers, one of the reasons being thenegative feedback used in its design. Thus, considering that the978-1-7281-6870-8/20/ 31.00 2020 IEEE127
transimpedance provides a fair tradeoff between gainbandwidth product and noise, this solution is widely used inindoor [14]-[15] and outdoor [6], [7], [16], [17] VLCapplications, being the most popular choice in this domain[18]. At the amplification stage, the signal is post-amplified forthe data and clock recovery block. The post-amplifier does notadd a significant noise in the system. Still, it is best tominimize the noise contributions anyway, especially thecommon-mode noise (from either the external electromagneticsignals or the switching power supply noise). One solution toaddress this issue is by choosing a differential amplifier for thepost-amplification stage [19].Considering the outdoor usage of the VLC system inautomotive applications, and taking the strong parasiticsunlight into account [20], a compensation block can be addedto attenuate the saturation when the receiver is subjected todirect sunlight.To address the most common issues, an instrumentationamplifier is chosen as a foundation for this work. In order toimprove the results, the instrumentation amplifier was modifiedwith offset cancellation loops, based on two integrator circuitsfor sensing the DC output offset, T-networks being added inthe feedback paths to reduce the effect of parasitic capacitances[21]. From this point, the output signal can be further processin the analog-digital conversion (ADC) stage until the datasignal is reconstructed.III.CONSIDERATIONS REGARDING THE PRACTICAL DESIGNOF THE TRANSIMPEDANCE CIRCUITThe development of the transimpedance circuit begun withthe aim of having a minimum of 1 mVpp output for an input of1 nApp, which means that a gain of around 1 MΩ is required.During the design stage, a classic transimpedance amplifier isconsidered as the starting point. Thus, the circuit illustrated inFig. 1 is considered as a model for bandwidth analysis. Theshunt resistance (RSH) has usually a very high value, so it canbe ignored. On the other hand, the input capacitance has a highimpact on stability, bandwidth and noise [22]. The total inputcapacitance (CS) is the sum of the photodiode capacitance (CD),the common-mode capacitance of the amplifier (Cicm) and thedifferential capacitance of the amplifier (Cid). Cicm and Cidinclude both the board layout and the operational amplifierparasitic capacitance in accordance with eq. 1.Fig. 1. Basic transimpedance circuit model.CS CD Cicm Cid128(1)A. Stability AnalysisFirst, the non-infinite open-loop gain for a single pole opamp model will be examined. The Bode plot illustrated in Fig.2 shows the close-loop gain superimposed over the open-loopcharacteristic of the operational amplifier. This gain begins at 0dB, and at the frequency fz (determined by RF and CS) isstarting to rise up. Uncompensated, the close-loop gain curvewill rise and cross the operational amplifier open-loop gain AOLat the frequency fi. In order to avoid instability, a pole can beintroduced at a frequency fp with a compensation capacitor CFin parallel with the feedback resistor RF, which will limit theclose-loop gain at 20log(1 CS/CF). Imposing the condition fp fi, the largest bandwidth will be obtained by lowering the CF atthe minimum value, which is given by eq. 2.CF CS2πRF GBP(2)In order to maximize the bandwidth, the solution is todecrease the value of CS and/or to increase the GBP. Loweringthe value of CS has the advantage of limiting the noise gain to asmaller value and to increase the zero response fz to higherfrequency. This means that an operational amplifier with lowinput capacitance and a photodiode with a low junctioncapacitance CD must be considered. One adequate choice is theDET10A photodetector as it has a capacitance of only 10 pF.On the other hand, the sensitive area of the photodiode is alsoan important parameter, which in this case has only 0.8 mm2.To increase the communication distance as required inautomotive applications, the DET32A photodetector seems amore adequate choice as it has a capacitance of around 40 pF,but a larger sensitive area of 13 mm2. Finally, because unitygain operational amplifier is not mandatory for atransimpedance application, a decompensated amplifier is agood choice, as it offers better voltage noise specifications andlarger gain-bandwidth products. The OPA858 operationalamplifier is an adequate choice in this case.Fig. 2. Bode plot of the loop gain for thetransimpedance operational amplifier configuration.
The gain is given by the relations 3 and 4.(3)VO –V – A(s )A ω(4)A(s ) OL As ωAUsing Laplace transform function, the gain for thetransimpedance operational amplifier is given by the relation 5.VO(5) IDAOL ωARF RF (CS CF ) ωA ( AOL 1)CF 1 s 2 s ωA 1 AOL R (C C )()C CRC CSFFSFFSF The Gain-Bandwidth Product (GBP) is given by eq. 6.AOL ω A(6) GBP2πFrom here, the further calculation will lead to this relation:VOAωO2 RF OL IDAOL 1 s 2 s ωO ω 2OQbackground light up to 15,000 µW/cm2. Nevertheless, the aimof this work is to further improve the transimpedance circuitwith better components, additional compensation circuits forbackground light and also with common-mode noise reduction.In Fig. 3, a topology which addresses the aforementionedproblems is presented. Here, it should be underlined thatintrinsic noise is at a higher level than that of a classical TIA.Even so, when coupled interference is predominant, the noiseis better addressed with a differential configuration [24]. U1and U3 are transimpedance amplifiers with resistive Tnetworks formed by R1-R2-R3 and R5-R6-R7 respectively,which can contribute to the increase of the bandwidth byreducing the effect of the parasitic capacitance. The outputs ofthese amplifiers are returned to the positive inputs through U2and U4, which are in a classic Miller integrator topology. Theirpurpose here is to compensate the DC output offset appeareddue to the sunlight or other external sources of light. Bothoutputs from the U1 and U3 are injected into a differentialamplifier U5, in order to reject the common-mode interferingsignals.(7)where:ωO ω A ( AOL 1) f (2π )RF (CS C F )ωA ( AOL 1)RF (CS CF )Q CF 1 ωA 1 AOL(C CRCSF FS CF ) Using the following algebraic simplifications:CS CF( AOL 1)wA AOL wA 2π·GBP CF CF 1 AOL AOLC CCFs F Cs these will lead to the eq. 13 and eq. 14.fi f z ·GBPfiQ f z fc(8)(9)(10)(11)(12)(13)(14)A. Bandwidth ConsiderationsAn adequate VLC receiver must be able to work in anextended range of irradiance. The problem is that in outdoorconditions, the background light can have an irradiance goingfrom a few µW/cm2 up to more than 100,000 µW/cm2. Thisparasitic light can heavily disturb the useful data. In theseconditions, if the transimpedance circuit is too sensitive, thiscan lead to saturation. In [23], it was demonstrated that alogarithmic transimpedance circuit can compensate theFig. 3. Proposed transimpedance schematic.129
IV.SIMULATIONS RESULTSBefore proceeding to the simulation section, someconsiderations are still required. As both branches of thedifferential amplifier are identical, only one of them isconsidered, while the results will be replicated for the other aswell. So, U2 has a transfer function given by eq. 15. 1(15)sR4C2This will return for U1 an output voltage over input currentas follow:H (s ) R feqVoU 1R3 R4C2 s ·ID(1 R1C3s ) R2 R3 R4C2 s(16)where: R (17)R feq R1 1 2 , when R1 R2 / R3R3 One can see that there are two poles at – 1 / R1C3 and atparasitic inductance and capacitance of the PCB traces can be afactor for instability at high frequencies.As for the lower cutoff frequency fL, it is enough to chooseits value based on fL 0.1fi 100 kHz to avoid signaldistortion. From eq. 19, it can be determined the time constantR4C2 5.4 µs. For R4 100 Ω and C2 56 nF, a fL 96.9 kHzvalue is obtained. The TINA simulation results showing thebode diagram are presented in Fig. 4.As all the elements for one branch are now available, thegain of this stage can be determined based on eq. 16 and 17.Thus, the gain of this stage is 66 kΩ when the diode SD1 is notconducting, meaning that an output of at least 66 µV will beobtained for the minimum input current. The differential inputfor U5 will be in that case 132 µV, which establishes thenecessary gain for this stage to around 8. As the values of R9,R10 are established at 470 Ω, whereas R11, R12 are equal to 3.9kΩ, a total gain of around 1 MΩ is obtained.– R2 / R3 R4C2 , and one zero at the origin. Then the lower 3 dBcutoff frequency fL and the upper 3 dB cutoff frequency fH aregiven by eq. 18 and 19.R2(18)fH 2πR3 R4C21(19)fL 2πR1C3The communication frequency fi is settled at 1 MHz whichis high enough for the majority of the commercial vehicularLEDs. As the DET32A’s photodiode has a rise time τr of 14 ns,significantly higher than that of an OPA858 (i.e. 0.3 ns), itshould be verified if this could be an issue in choosing theproper fH. Thus, by applying the following relation 20 [25]:0.35(20)f3dBit is easy to determine that the chosen photodiode is not alimiting factor until f3dB 20 MHz, and the margin is safeenough for a frequency fH 10 MHz. It is best to have enoughgain on this stage, but one must pay attention to the amplifier’ssaturation by ambient light. One can prevent this issue byinserting a diode SD1 in parallel with R1 and choosing R2-R3accordingly. 1PS76SB17 is an ultra-high-speed switchingSchottky diode with a very low capacitance (0.8 pF) and a verylow forward voltage (0.3 V) suitable in this case. Theoperational amplifier has a 2.5 Vpp output swing, so R2 22 Ωand R3 75 Ω can be used. In order to have a reasonable valuefor C3 and a gain of around 50 kΩ for this stage, R1 is settled at15 kΩ. From eq. 18, one can now determine C3 1/2πR1fH 1/(2πR1·10 MHz), which is around 1.06 pF, the maximum valuefor the chosen band. Taken eq. 2 into account, the stabilitycondition imposes that C3 must be greater or equal to 0.13 pF.A value of 1 pF is satisfying both conditions and it is settingthe fH to 10.6 MHz. As it is very much possible that theparasitic capacitance in the feedback path to be around thisvalue, precaution must be taken in designing the PCB, becauseτr 130Fig. 4. Bode diagram simulation results.V.DISCUSSION ON THE NOISE PERFORMANCESThe main sources of internal noise in a photodiodeamplifier are shot noise, thermal noise, dark noise, flicker noiseand amplifiers’ current and voltage noises. The highest impacton the signal-to-noise ratio will be from the first stage, so
initially U1-U2 circuit will be considered using the noisemodel from Fig. 5 [26].For a photodiode, the dominant source of noise isdetermined by the statistical uncertainty due to the discreetnature of the light and of the electrical current, called shotnoise. This type of noise is usually found in semiconductorjunctions, where the electrons pass the potential barrier in adiscreet manner [24]. For a given photocurrent IPHD, the noisecontribution is given by eq. 21.I shot 2qI PHD Δf(21)For resistors, the dominant source of noise is the NyquistJohnson noise, also called thermal noise. The contribution inthis case is given by the formula:Ethermal 4kTRΔf(22)For the FET operational amplifiers, the input current noisescontribution can be ignored, as the main source of noise beingthe voltage ones. Apart from the noise gain, the signal gain isalso required to estimate the SNR [19].The amplifier’s signal gain for U1 is determined from eq.17. From the datasheet of the amplifier, it is easy to see that theflicker noise is relevant up until around 100 kHz and it isnegligible in our frequency domain of interest (i.e. 1 MHz).The Schottky diode will generate a lot less noise than theresistors in the feedback path, so it can be ignored as well.2 R E 2 4kTR3 Δf 32 nV 2 R3 2E R2out 4kTR2 Δf 12 nV 2(24)2R3out(25)kT(26) 0.0016 nV 2π 2 R4 C 22One can see that the dominant source of noise here is thethermal noise of the feedback resistor. The noise of thephotodiode will be amplified by the signal gain, so this cannotbe improved. Regarding noises added by the integratoroperational amplifier, these are low enough to be also ignored.2EintR4 Finally, en1 will be amplified by the noise gain, which isdetermined by the following relation 27 [27].enoe R2 1 R feq (C3 C D )s 1 eni R3 1 R feq C3 s(27)It can be seen that there is one zero at 1 / R feq (C3 C D ) ,and one pole at 1 / R feqC3 . So, we have f pnoise 1 / 2πR feq C3and f znoise 1 / 2πR feq (C3 C D ) , which is the same as fH.The output noise is frequency dependent, with five distinctregions: region 1, dominated by flicker noise, region 2, with adirect transfer of the noise to the output, interrupted by the gainpicking in region 3, followed by a plateau in region 4, and byan AOL roll-off to infinite frequency [19]. Typically, this outputnoise component is dominated by the region 4 and 5. In thiscase, these values are given by eq. 28 and 29 leading in the endto relations 30 and 31.2E noe4 (1 C D )2C23 C3GBP 21 ·eni C C2πR feq C3 D 32E noe5 GBPC3 C D 2eniC3(28)(29)22Enoe4 1,350 nV(30)22Enoe5 1,400 nV(31)For the final output noise, the result is given by eq. 32:E noe 2R11 222E R1out E noe4 E noe 5 230 nVR9()(32)At the minimum input signal, Vout 1 nA x 1 MΩ 1 mV,the signal to noise ratio is given by 33.Figure 5. Noise analysis circuit.From Fig. 5, there are the following noise contributors atthe output:2E2R1out R 1 2 4kTR1Δf 4,000 nV 2 R3 (23)Vout(33) 72 dBEnoeThe simulations provide encouraging evidence that theproposed circuit is able to cope with a parasitic light going upto 100,000 µW/cm2 when the useful data signal is generating aphotodiode current going up to 200 nA. Thus, this is a goodSNR[dB ] 20log131
indicator that noise resilience can still be enhanced,contributing this way to vehicular communications compatibleVLC systems. These results showed that this schematic isadequate for visible light communication in outdoor conditions,being able to cope with strong sunlight, to offer a goodprotection to noises and interferences, and with a high stability.On the downside, it should be mentioned that if it is to comparethe proposed schematic with other proposals, this one has ahigher complexity, which imposes a careful design for the PCBin order to eliminate the instability at high frequencies.Additionally, for an optimum common-mode performance, thecircuit requires precision passive components’ values. Hence,the overall cost is estimated to be higher.VI.CONCLUSIONSThis paper provided an analytical evaluation based onsimulations for a photodiode amplifier intended for VLCtechnology in automotive applications. 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Photodiode Amplifier with Transimpedance and Differential Stages for Automotive Visible Light Applications Cătălin Beguni1,2, Alin-Mihai Căilean1,2,, Sebastian-Andrei Avătămăniței1,2 and Mihai Dimian1,2 1Department of Computers, Electronics and Automation, Stefa
The MAX3657 transimpedance amplifier is designed for 155Mbps fiber-optic applications. The functional dia-gram of the MAX3657 comprises a transimpedance amplifier, a voltage amplifier, a DC-cancellation circuit, and a CML output buffer. Transimpedance Amplifier The signal current at the input flows into th
Oct 12, 2016 · The signal path consists of a transimpedance amplifier stage, an AGC voltage amplifier, and a CML output buffer. The on-chip filter circuit provides a filtered VCC for the photodiode and for the transimpedance amplifier. The DC input bias circuit and automatic gain control use
corner frequency. The second stage of the transimpedance amplifier is a differential amplifier which was previously described. Figure 5 A depiction of the photodiode and the original transimpedance amplifier design. So why was the differential topology chosen? Originally we had a different de
transimpedance gain, input referred noise and group delay variation. A 100 Gb/s transimpedance amplifier (TIA) for optical receiver front end is designed in this work. To achieve wide bandwidth and low group delay variation a differential TIA with active feedback network is proposed. P
E80 Lecture 4.2: Basic Electrical Measurements Agenda: Operational Amplifier o Recap: Non-inverting amplifier and unity gain buffer o Inverting amplifier (multiplication) o Summing amplifier (add and subtract) o Differentiator and integrator o Difference amplifier o Instrumentation amplifier o Transimpedance amplifier o Active filters 2
Small-Signal Transimpedance Z21 MAX40659, IIN 1µAP-P 36.4 kΩ OFFSET Input Transimpedance MAX40658 4.7 kΩ MAX40659 9.4 Input Clamping Current 100 mA Output Common-Mode Voltage MAX40658 VCC - 0.125 V MAX40659 VCC - 0.25 Differential Output Offset ΔVOUT IIN 0
1.1 Generic Fiber Optic Communication System Figure 1.1 depicts a typical interface of an optical communication system [1-3]. It consists of both digital (MUX and DEMUX) and analog building blocks such as a transimpedance amplifier (TIA), a limiting amplifier (LA), laser driver, clock and data recovery (CDR), and other sub components.
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