Flyback Transformer Design Considerations For Efficiency .
Flyback transformer designconsiderations for efficiencyand EMIBernard KeoghSystem and Solutions EngineerHigh Voltage Power SystemsIsaac CohenPrincipal System ArchitectLow Power Controller and ConverterTexas Instruments
Power Supply Design Seminar 2016/17AC/DC power supplies widely use the flybackconverter given its simplicity and wide operatingrange, and because it eliminates the output filterinductor and free-wheeling rectifier required forforward-mode topologies.Three main topology components dominate flyback-converter performance: theprimary switch, secondary rectifier and transformer. This paper focuses on theimportance of transformer design, since this single component has a profound impacton converter efficiency and electromagnetic interference (EMI). This paper will discussthe various conflicting design requirements, the often-neglected subtleties of core lossand snubber clamp level, and ways to improve transformer performance.Introductionoften-neglected absorption of magnetizing energyMany AC/DC and DC/DC power supplies, from veryby the snubber is also highlighted.low power levels to as much as 150 W or more,For conducted EMI, we will outline the causesuse the flyback converter. Often maligned and notof common-mode (CM) EMI, suggesting variousalways fully understood, the transformer is thewinding structures and techniques to ensure goodheart of the flyback power supply and probably theCM balance.most important component. When designed andFinally, through several examples we will show howimplemented well, the transformer can deliver thetransformer construction can have a significantrequired performance cost-effectively. When poorlyimpact on both efficiency and conducted EMI. Indesigned, it can cause EMI issues, low efficiencythese examples, we changed none of the otherand possible thermal overstress issues.components – only the transformer – in order toThis paper will discuss the causes of the majordemonstrate how a well-designed transformer canlosses in the flyback transformer. In particular, wesimultaneously improve both efficiency and EMI.will review core loss in light of recent researchfindings that highlight the significant impact of dutyThe flyback topologycycle and DC bias. The significant contributionThe flyback transformer is not really a transformerof proximity effect on AC copper loss is alsoin the conventional sense; it is actually a coupleddiscussed.inductor. Figure 1 is a simplified schematic of aWe will review wire-size selection and windingmethods to reduce AC copper loss. The effect ofsnubber clamp voltage levels and theTexas Instrumentsflyback converter. The flyback transformer in thisexample has three windings: primary, secondaryand bias (sometimes called the auxiliary winding).2September 2016
Power Supply Design Seminar 2016/17When the primary switch turns on, the input voltagerequired by the pulse-width-modulation (PWM)is imposed across the primary winding. Since thecontroller for regulation, the primary switch isdot-end of the primary winding is connected toturned off. The primary current then transfers to theground, the dot-end of both the secondary andsecondary winding and the current decays at a rateauxiliary windings will be negative and proportionalproportional to VOUT. In this way, the energy stored into the input voltage. The respective rectifier diodesthe transformer during the buildup of primary currenton those windings will thus be reverse-biased.gets released to the load and output capacitorWhile the primary switch remains on, current buildsduring the flow of secondary current. This is, ofup in the primary winding at a rate dependent oncourse, a simplified explanation; for more detailedthe input voltage and the primary magnetizingdescriptions of the flyback topology and modes ofinductance, LP.operation, see references [1], [2] and [3].Based on this description, the flyback transformeractually operates as a coupled inductor, whereBIASSECVOUTcurrent builds up to a peak value in the primarywinding and then decays back down in thePRIMsecondary winding during the flyback interval. VINThus, when designing the flyback transformer andassessing the losses, you must consider it more ofan inductor than a transformer.DRVVDDCSPWMControllerFlyback operationFigure 2 shows the different operating phasesFBof the flyback converter during a single switchingFigure 1. Simplified schematic for a typical flyback converter.cycle, with the corresponding voltages and currentsshown in Figure 3. During the primary switchOnce the current in the primary reaches the levelVIN PRIMSECon-time interval in Figure 2a, current flows fromVOUTVIN PRIM(a)VINPRIMVOUTOFFON SECSEC(c)VOUTVIN PRIMSECVOUTOFFOFF(b)(d)Figure 2. Flyback converter operating intervals per switching cycle: primary switch on-time (a); primary switch turn-off, transition interval (b);secondary rectifier clamping and conduction interval (flyback interval) (c); discontinuous conduction mode (DCM) ringing interval (d).Texas Instruments3September 2016
Power Supply Design Seminar 2016/17(a)(c)(b)(d)Figure 3. “Idealized” flyback converter voltages and currents, with highlighted operating intervals per switching cycle: primary switch on-time (a);primary switch turn-off, transition interval (b); secondary-rectifier clamping and conduction interval (flyback interval) (c); DCM ringing interval (d).the input-voltage source through the transformer’scapacitance on the switch node. The losses in themagnetizing inductance, storing energy in thetransformer core and the AC resistance (ACR) of theinductor air gap. During the transition interval inwindings dampen this ringing.Figure 2b, the primary current transitions to theIn continuous conduction mode (CCM), the intervalsecondary, while the transformer’s primary voltagein Figure 2d does not occur because the primaryswings positive. When the transformer primaryon-time commences before the secondary currentvoltage swings sufficiently more positive than VIN,decays to zero. In CCM, not all of the energythe output flyback diode becomes forward-biasedstored in the transformer’s magnetizing inductanceand clamps the voltage. Subsequently, duringtransfers to the secondary during each switchingthe interval in Figure 2c, the secondary currentcycle.will decay linearly (since the voltage across thesecondary winding is negative). During the intervalFlyback transformer lossesin Figure 2c, some or all of the energy previouslyThe flyback transformer is responsible for a largestored in the transformer’s magnetizing inductancepercentage of the total losses in a flyback powerwill be released to the secondary-side storagestage. There are four categories of losses:capacitor and to the load. Core losses.In discontinuous conduction mode (DCM), all of the Copper (winding) losses.energy stored in the inductance during the primary Transition losses.on-time interval is delivered to the secondary duringthe flyback interval. In this mode, the secondarycurrent decays to zero at the end of the flybackinterval. Subsequently, the interval in Figure 2d isthe DCM ringing interval, where the magnetizinginductance resonates with the total parasiticTexas Instruments External losses.Core losses occur in the transformer’s ferrite coreand depend on the core’s flux density (amplitude,duty cycle and flux-density rate of change),frequency of operation, core size or volume, and4September 2016
Power Supply Design Seminar 2016/17properties of the chosen ferrite material. Differentare significant causes of loss. As transition loss ismaterials optimized for different frequency and peakbeyond the scope of this topic, see reference [4] forflux-density ranges will exhibit varying core-lossfurther details.characteristics. We will describe core losses in moreWhile the transformer itself incurs most of thedetail in the next section.losses, two significant external losses occur due toThe flow of current through the resistance of theparasitic elements of the transformer. First, leakagewindings causes copper or winding losses. Mostinductance results in a loss incurred in the externaldesigners refer to it as copper loss because copperclamp or snubber circuit, which is necessary tois by far the most commonly used wire materialkeep the voltage stress on the primary switchgiven its low resistance, ease of manufacture andbelow its VDS maximum rating. Second, transformerwide availability.capacitance contributes to the total parasiticCopper loss breaks down further into DC loss andcapacitance of the switch node. An increase in theAC loss. DC loss is caused by DC or low-frequencyswitching node capacitance increases the switchingroot-mean-square (rms) current flowing through thelosses in the primary switch. We discuss the effectsDC resistance (DCR) of the winding. Maximizing theof leakage inductance and interwinding capacitancewire cross-sectional area and minimizing the wirefurther in the section on EMI shielding.length minimizes DCR.AC loss is caused by high-frequencyelectromagnetic effects from the magnetic fieldproduced by the time-variant current flowing in theCore loss in a flyback converterTraditionally, designers assumed that DC flux didnot affect the core losses in an inductor, and theselosses are largely independent of the flux-densitywires. AC loss can be very significant, especially forwaveform. For example, as Figure 4 shows, thelarge wire diameters. We will discuss AC losses inflyback flux-density waveform is nonsinusoidal, notmore detail later.necessarily 50 percent duty cycle and contains aTransition losses refer to the losses associatedsignificant DC component. Yet when calculatingwith the transition or commutation of transformercore loss, most designers neglect the DCcurrent from the primary to secondary winding.component and duty cycle and consider only theIn this region, the rate of change of the currentspeak-to-peak flux swing, as shown in Figure 5.(di/dt) is very high, so the currents will have largeAnother common assumption is that all waveforms,high-frequency harmonic content. Also in thisregardless of duty cycle and DC bias, have theregion, since both primary and secondary currentssame core loss because they have the same Bpk-pkflow simultaneously, the flyback transformerflux-density swing. Thus, designers extracted thebehaves more like a conventional high-frequencycore losses from the published sine-wave-specifictransformer, and so high-frequency effects and ACRloss curves using the flux-density amplitude andfrequency experienced by the converter.Texas Instruments5September 2016
Power Supply Design Seminar 2016/17offer equations that relate manufacturer-publishedFlyback Waveforms at CCM/DCM Boundaryspecific loss data for sine-wave excitation only (noVINΔBacBpk-pkBDCDC bias) to actual losses generated with rectangularwaveforms and DC bias, and provide empiricalsupport for their theories.VREFLECTEDTSW 1/FSWFigure 4. Flyback transformer flux-density waveform at CCM/DCMboundary.Flyback Waveforms, Neglect BDC and DEffect of rectangular waveforms withvariable duty cycleReference [6] investigates the ratio of core lossunder rectangular-wave excitation to that of asinusoidal-wave excitation of equal flux amplitudefor a number of magnetic materials (Figure 6ΔBac Bpk-pkreproduced from [6]). It also introduces a curvefit equation for the core loss versus duty cycleTSW 1/FSWFigure 5. Flyback transformer waveforms at CCM/DCM boundary,neglecting the BDC component and duty-cycle variation.A closer qualitative examination reveals that theseassumptions must be incorrect. It should beapparent that the eddy currents induced in the coreare higher when the rate of flux change is faster,since the induced voltage driving the eddy currentswill be higher. Thus, compared to the eddy current(Equation 1): (1)where D is the duty cycle and γ is a correction factorspecific to the material, operating frequency andtemperature, and has to be extracted from carefulmeasurements. Reference [6] tabulates measuredvalues of γ for several different ferrite materials.loss generated by a sine wave of equal frequency,a low duty-cycle rectangular voltage waveformgenerating an equal peak-to-peak flux density mustgenerate higher eddy current loss in the core.PV RECTPV SINAdditionally, the magnetic domains theory suggeststhat the domain walls cause nonuniform flux density,which results in eddy current losses in excess ofthose related to the material’s conductivity.References [5], [6], [7] and [8] discuss in moredetail the mechanisms that relate the losses towaveforms, duty cycle and DC bias, which areFigure 6. Core-loss ratio for rectangular versus sinusoidal excitationas a function of duty cycle [6]. (Image: Courtesy of the Institute ofElectrical and Electronics Engineers [IEEE], IEEE 2014)beyond the scope of this paper. Those authorsTexas Instruments6September 2016
Power Supply Design Seminar 2016/173F35 at 500 kHz (various Bpk, D values)Curve fit: F(HDC) 2.1875 x 10-4 (HDC)2 1PC90 at 1 MHz (various Bpk, D values)Curve fit: F(HDC) (0.04 x HDC 1)0.5Figure 7. Core-loss ratio for DC bias versus no bias excitation. Source: Reference 5. (Images: Courtesy of Virginia Tech)Effect of DC biaswhere PV SINE is the Steinmetz equation loss forThe author of reference [5] measured the effectsinusoidal excitation.on core losses when adding a DC bias, HDC, forReferences [5] and [6] contain the informationrectangular-waveform excitation and proposednecessary to use Equation 2 for several Ferroxcubea curve-fitting factor, F(HDC), to account for thematerials. We hope that magnetic materialsincrease in loss due to the presence of DC bias inmanufacturers will consider verifying the validity ofthe core.the results reported and generate the informationFigure 7 shows curve-fitting equations for F(HDC)necessary to enable users to accurately calculatefor 3F35 and PC90 ferrite materials generated fromcore losses in PWM applications, which are farmeasurement data of core losses with DC bias, andmore common than sine-wave applications.rectangular-waveform excitation at different dutyIn order to put Equation 2 to practical use,cycles and flux-density amplitudes.manufacturers must make available the followingThe F(HDC) function represents the increase in coreinformation about magnetic materials:loss caused by DC bias; it appears that it is relatively Frequency and flux-density exponents to generateinsensitive to the amplitude and duty cycle of thethe correct PV SINE at the relevant flux density andexcitation voltage.frequency range. (Note: Ferroxcube provides an excellentTotal core loss for arbitrarywaveformsEquation 2 combines the results presentedabove with the core-loss equations provided bymanufacturers for sine-wave excitation to calculatespreadsheet documenting their materials, availableupon request). The γ parameter and an appropriate equation with whichto use it. The equation for FDC(HDC ).core loss for the rectangular-waveform excitationpresent in flyback (and many other PWM) converters:𝑃𝑃! !"!# 𝑃𝑃! !"# 𝐹𝐹!"# %&'( (𝛾𝛾, 𝐷𝐷) 𝐹𝐹!" (𝐻𝐻!" )(2)Texas Instruments7September 2016
Power Supply Design Seminar 2016/17We must emphasize a few points: Duty cycle and DC-bias effects on core losses aresignificant, and should not be ignored. The substantial increase in core loss at extreme dutycycle values is an often-neglected penalty of wide input/output-voltage-range converters. The increase in loss due to DC bias reduces the benefitswill depend on the frequency, wire diameter andoverall layer structure. The eddy currents inducedinside the wires (as a result of the magnetic fieldinside the wires) are the main cause of AC loss andincreased ACR. These eddy currents lead to skinUseoriginal Excel file foreffect and proximity effect, which we will explaina better qualty image.further in the next sections.expected from CCM operation.Current Harmonic Content1.2 The assumption of equal core loss in single- and doubleended applications with equal AC flux excursions isCopper loss and AC effects inflyback transformersCurrent flowing through the resistance of copperwindings causes copper loss in a transformer.16.05.0Harmonic Current (A)probably incorrect.7.0IPRI 4 A peakNp/Ns 5DPRI 20%DSEC 35%0.84.00.63.00.42.0IPRIISEC0.21.0Losses arise because of the DC component of0the current and the DCR of the windings, but also024For flyback converters operating in DCM or81012141618200.0Harmonic Order(and often more significantly) from high-frequencyAC effects.6Figure 8. Harmonic content of typical flyback primary and secondarycurrent.transition mode (TM), the current flowing in bothSkin effectthe primary and secondary windings is triangularWhen DC current flows in a wire, the current densityin shape (Figure 3). Since the flyback converteris uniform throughout the wire’s cross-section; instores energy during the primary conduction intervalother words, the current is distributed equally acrossand then delivers energy to the load and the outputthe wire. But when a time-varying AC current flows,capacitor during the secondary conduction interval,the changing current produces a changing magneticthe duty cycle of each interval is typically lessfield around the wire. This changing magnetic fieldthan 50 percent. The primary duty cycle will oftenis also present inside the wire. Faraday’s law statesbe much less than 50 percent at high-line inputthat whenever there is a changing magnetic field,voltages, where the di/dt of the current ramp isa voltage (or electromotive force [EMF]) is induced,much steeper, and so the high-frequency harmonicso as to oppose the changing magnetic field.content will also be greater. Consequently, AC lossThe induced voltage causes circulating eddymechanisms can become more significant. Notecurrents to flow, and since the conductivity ofthat in Figure 8, the zeroth harmonic is actually thecopper is high, these currents can be veryDC component of the current waveform.significant. The eddy currents reduce or cancel theSince the flyback’s primary and secondary currentshave a significant DC component and significanthigh-frequency harmonic content (Figure 8), bothACR and DCR are important. The ACR-to-DCR ratioTexas Instrumentscurrent flow in the center of the wire, and reinforceor increase the current flow in the outer regions ofthe wire cross-section, leading to current-densitydistribution as shown in Figure 9.8September 2016
Use original Excel file fora better qualty image.Power Supply Design Seminar 2016/17Skin Depth vs. Frequency0.8Current density, J1/e0.710.6Depth (mm)07d5d0.50.40.30.20.1Figure 9. Nonuniform AC current distribution due to induced eddycurrents. Source: Reference 9.0.00200400600800Frequency (kHz)As the frequency of the AC current increases, thecurrent becomes more concentrated near theFigure 10. Copper skin depth (or penetration depth) in millimeters vs.frequency in kilohertz at 100 C.outer edges of the wire, and the central portionACR to DCR, assume that all of the AC currentof the wire will carry almost none of the current.flows in an annular ring around the outside of theThe “skin depth” is defined as the depth insidewire, one penetration-depth wide. Thus, Equation 5the wire where the current density has fallen toapproximates the ratio of ACR to DCR by the ratioapproximately 37 percent (1 e) of the value at theof the total wire cross-section to the cross-sectionsurface. This depth is also where the penetratingof the 1-δ wide outer annulus:magnetic field strength has fallen bythe same 1 eratio – hence it is also sometimes referred to as“penetration depth.” Penetration depth, δ, dependson the resistivity of the wire material, ρ, the relativemagnetic permeability of the wire material, μr, andthe frequency of interest, f. See Equation 3:𝛿𝛿 𝜌𝜌(𝜋𝜋 𝜇𝜇! 𝜇𝜇! 𝑓𝑓)This illustrates the significance of skin effect whenusing large diameter wires. Using the examplediameter to 2-δ reduces the ACR to DCR ratio(3)to approximately 1; however, the DCR will haveincreased twelvefold due to the significantly smallerwire diameter. Filling the space occupied by theexclusively copper, δ can be conveniently expressedas a function of only frequency. At 100 C, Equation 4gives the δ of copper, where f is in kilohertz (plottedin Figure 10):𝛿𝛿 (5)from Figure 9 and Equation 5, reducing the wireSince the wire used in transformers is almost2.3 10!!12.4 𝑚𝑚𝑚𝑚 !!(𝜋𝜋 4𝜋𝜋 10 1 1𝑘𝑘) )!4949 2!!𝐷𝐷𝐷𝐷𝐷𝐷 (7𝛿𝛿) (5𝛿𝛿)49 25 24single 7-δ wire with multiple 2-δ wires reducesDCR and consequently ACR. Replacing the singlelarge 7-δ wire with an array of nine paralleled 2-δwires (to fit in approximately the same total area asthe original wire), DCR is now 136 percent of theoriginal value (72/(9*22)). Thus ACR is now 1.36 times(4)the original DCR, compared to twice the originalDCR for the single large-diameter wire. Of course,Looking back at Figure 9 as an example, thethis improvement comes at the penalty of morewire diameter is seven times larger than δ at thecomplicated multistranded wires – but these are kindfrequency of interest. To approximate the ratio ofTexas Instrumentsof trade-offs that you need to consider when weighingcost/complexity against efficiency performance.9September 2016
Power Supply Design Seminar 2016/17(a)(b)(c)Figure 11. AC current distribution due to induced eddy currents for single wire (skin-effect only) (a); two adjacent wires with current in samedirection (b); and two adjacent wires with current in opposite directions (c). Source: Reference 9.Proximity effect – single layerWhen current flows in two adjacent wires, theIn the previous section, we explained skin effect in themagnetic field from the AC current flow in each wirecontext of a single isolated wire. But rarely will youaffects the current distribution of the other.encounter a single isolated wire in practice. Flyback-When currents flow in the same direction, thetransformer windings always consist of multiplecurrent distribution will tend toward the farther-turns, built up in multiple layers, including at leastaway outer surfaces, and the current density at theone primary winding and one secondary winding.facing edges drops. When currents flow in oppositeThey usually also include an auxiliary winding, anddirections, the current density concentrates at thesometimes multiple secondary windings.inner-facing surfaces.Skin effect alone is actually not that significant. Whatis far more important in the context of transformersis “proximity effect.” This is very similar to skin effect,but arises from the effect of the magnetic field thatAC current flow in one wire causes on all adjacentwires. As you will see, proximity effect can build uprapidly as you add more layers of wire – to the pointwhere the inner layers are carrying significantly moreeddy current than load current.We will first explain how proximity effect occurs ina pair of wires and then in a single layer of multiplewires. A common misconception is that proximityeffect only applies to multiple-layer windings and doesnot occur in single-layer windings. But proximity effectdoes occur in single-layer windings, and its extentdepends on the chosen wire diameter.If you place multiple adjacent wires together in atypical single-layer transformer winding, the currentflow will be in the same direction in each wire,assuming that they are connected in series. Theproximity effect will reduce the current density at theadjacent-facing edges of each wire (except for thefirst and last wire in the layer), as shown in Figure 12.The current density is concentrated along thetop and bottom surfaces of the wires in the layer,with little current flow in a central strip alongthe layer. This qualitatively highlights how muchmore significant and important proximity effect iscompared to skin effect alone. Even for a singlelayer, if the wire diameter is too large compared tothe penetration depth, proximity effect will occur.Figure 11 shows the AC current flow in a singlewire, the effect of two adjacent wires with currentflow in the same direction, and the effect of currentsin opposite directions. Note that for ease ofFigure 12. AC current distribution due to proximity effect for asingle-layer winding, with all currents flowing in the same direction.illustration, the wire diameter is much greater thanthe penetration depth at the frequency of interest.Texas Instruments10September 2016
Power Supply Design Seminar 2016/17adjacent layers and how the resulting currentconcentration is worse than skin effect alone.Proximity effect becomes progressively worsewith the addition of more winding layers, inducingFigure 13. AC current distribution due to proximity effect for atwo-layer winding, with all currents flowing in the same direction.canceling eddy currents in each layer thatcontribute significantly higher losses. Figure 15illustrates a three-layer 24-turn winding, with eight turns per layer. Current is flowing in the samedirection (out of the page surface) in each windinglayer. Once again, the wire diameter is much largerthan the penetration depth in order to highlight theFigure 14. AC current distribution due to proximity effect for a two-layer winding, with currents in each layer flowing in oppositedirections.proximity effect.winding over two layers, the proximity effect will If you extend and implement the transformer Proximity effect – multiple layers L3L2L1impact the distribution within each layer as alreadyseen – but each layer will also impact the other.Figure 13 illustrates how the current distributionFigure 15. AC current proximity effect for a three-layer winding, withcurrents in each layer flowing in the same direction.is concentrated only along the outer surface of theAssuming a normalized 1-A current in the winding,wires in each layer. A two-layer flyback transformerwith 24 turns the magnetomotive force (MMF) isprimary or secondary winding could typically have24 At. Since the wires are so large compared tothis kind of winding structure.the penetration depth, the magnetic field cannotArranging a two-layer winding with currents flowingpenetrate far enough into any of the winding layers.in opposite directions in each layer causes thecurrent density to concentrate along the innerfacing surfaces of the wires in each layer, asshown in Figure 14. A forward-mode transformerA corresponding 24-At MMF on the inner surface ofthe first innermost winding layer (L1) cancels the 24At of MMF of the air gap. Thus, the inner surface ofeach wire in layer L1 must carry 3 A each in order towould typically have this type of winding structure,generate 24 At of MMF across eight turns.where both primary and secondary current flowsSince they are all connected in series, the netsimultaneously in opposite directions. A flybackcurrent in each wire must be 1 A. This means thattransformer with adjacent primary and secondarya canceling 2-A current must flow in the oppositewinding layers has this type of structure duringdirection on the outer faces of the wires in L1 inthe transition interval, when the primary currentorder to get 1 A net. The magnetic field from thatcommutates to the secondary.opposing 2-A current on the outer face of L1 willThe illustrations in Figure 13 and Figure 14 arethen force a canceling 2-A current to flow in theof course grossly simplified, with very large wirediameters, to illustrate proximity effect betweenTexas Instrumentsopposite direction on the inner face of L2 as shownin Figure 15.11September 2016
Power Supply Design Seminar 2016/17Once again, since the net current in each wire in L2Proximity effect – passive layersmust be 1 A, yet another 1-A canceling current willPassive layers are layers of a winding structure thatflow on the outer faces of L2. The magnetic fielddo not carry any useful load current. In some casesfrom the 1-A current on the outer face of L2 forcesthey never carry useful current (such as an EMIa corresponding canceling current in the inner faceshield), while in other cases they carry current onlyof L3. Because the wire diameter is so large that thepart of the time (such as a center-tapped secondarymagnetic field cannot penetrate far enough into thein a forward-mode push-pull converter – eachwire, these canceling currents develop to allow thehalf only carries current 50 percent of the cycle atmagnetic field to propagate through the multilayermost). During any interval when no load currentwinding structure.flows, proximity effect-induced eddy currents canIn the example in Figure 15, the initial expectationis that the conduction loss would be proportionalto (3 * I2), since each of the three layers carries thesame net current, I. Using Equation 6 to sum thecontribution of the currents on all of the faces resultsin a loss proportional to:𝑃𝑃!"# 𝐼𝐼 ! 𝐼𝐼 ! (2𝐼𝐼)! (2𝐼𝐼)! (3𝐼𝐼)! 19𝐼𝐼 !(6)The total losses are more than six times higher thanflow in the nonconducting winding, contributing toconduction loss even when not conducting.A flyback transformer with interleaved primary andsecondary layers is another example of a passivelayer, when the nonconducting secondary issandwiched between conducting primary layers.Even with a noninterleaved flyback transformer, theprimary or secondary layer that sits closest to thecore air gap will also be a passive layer when it isnot conducting.expected. Adding more layers with the same largeFigure 16 illustrates this passive-layer proximitywire diameter makes the situation progressivelyeffect, where a single secondary layer, S, isworse. For four layers, the loss would be (44 * I2)sandwiched between two inner primary layersvs. (4 * I2), 11 times higher. For five layers, the loss(P1, P2) and two outer primary layers (P3, P4). Aswould be (85 * I2) versus (5 * I2), 17 times worse; andbefore, assume that the wire diameter is muchso on for more layers.larger than the penetration depth and that aIntuitively, you can see th
winding and then decays back down in the secondary winding during the flyback interval. Thus, when designing the flyback transformer and assessing the losses, you must consider it more of an inductor than a transformer. Flyback operation Figure 2 shows the different operating phases of the fly
Analysis and Design of Multioutput Flyback Converter A study For A Lab Upgrade on the Flyback converter assignment at Chalmers Elteknik Master's thesis in Electric Power Engineering . The result of the transformer design shows that a new assignment that can demonstrate how magnetic core behaves can be introduced. Keywords: Flyback, Multi .
A representative flyback converter can be seen in Figure 3. Figure 3. A Simplified Schematic of a Flyback Converter CONTROL Vin T Q Ipri Vout D The power switch essentially places the primary inductance of the flyback transformer across the input voltage source when it is turned
3. Instrument transformer: Used in relay and protection purpose in different instruments in industries . . Current transformer (CT) . Potential transformer (PT) . Open circuit and Short circuit Test on transformer . These two transformer tests are performed to find the parameters of equivalent circuit of transformer and losses of the transformer.
the inductor split to form a transformer. Figure2:- Operation of Flyback converter during ON state When the switch is closed shown in Figure 2, the primary of the transformer is directly connected to the input voltage source. The primary current and magnetic flux in the transformer increases, storing energy in the transformer. The voltage .
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Design of Efficient Flyback Switching Power Supply Based on LM5021 HOU Xingang1, a,WANG Lei1, b，CHEN Heng1, c，and CHEN Jinglei1, d 1 Xijing U nivers ty ,' aS hx , C 7 10 23 a 707861498@qq.com, b774383203@qq.com, cchenhenrys@qq.com, d 843699667@qq.com Keywords: flyback switching power supply; LM5021; current-limiting circuit; transformer Abstract. The paper introduces the process of .
Transformer Design & Design Parameters - Ronnie Minhaz, P.Eng. Transformer Consulting Services Inc. Power Transmission Distribution Transformer Consulting Services Inc. Generator Step-Up Auto-transformer Step-down pads transformer transformer 115/10 or 20 kV 500/230 230/13.8 132 345/161 161 161 230/115 132 230 230/132 115 345 69 500 34 GENERATION TRANSMISSION SUB-TRANSMISSION DISTRIBUTION .
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