A Compact X-Band Coplanar Waveguide Hybrid Lowpass Filter
International Journal of Electrical and Electronic Engineering & Telecommunications Vol. 9, No. 2, March 2020A Compact X-Band Coplanar Waveguide HybridLowpass FilterHamza O. Issa, Walaa S. Diab, and Mohammed W. WehbiDepartment of Electrical and Computer Engineering, University of Beirut Arab University, Debbeyeh, LebanonEmail: h.issa@bau.edu.lb; {walaa.diab; mohammed.wehbi}@live.comcan add further circuit complexity. Researchers using thistechnique reported miniaturization percentages around50%.Thirdly, the hybrid approach is heavily used due to itssimplicity in design. It is achieved by loadingtransmission lines with lumped elements (mainlycapacitors) [8]-[15]. It is usually referred to as the “semilumped” approach. The advantage of employing thesemi-lumped approach, compared to the use ofdistributed approach (only transmission lines), is clearlythe miniaturization. On the other hand, its advantagecompared to the purely lumped approach (exclusivelycapacitors and inductors), is to overpass the normalizedvalue problem. For these reasons, the semi-lumpedapproach is used in this work.Authors in [8]-[9] reported that by adding lumpedcapacitors to a transmission line while increasing itscharacteristic impedance permits to reduce the physicallength of the transmission line and thus overall devicedimensions. The capacitors can be added either at themiddle of the transmission lines [8] or at the extremities[9]. Inductors can be used instead of capacitors, but theysuffer from very poor quality factor especially at highoperating frequency [9].The use of localized surface mount components (SMC)for miniaturization has been proved very effective forfrequencies reaching no higher than 2-3 GHz. However,to the authors’ knowledge, no studies on the possibility ofusing localized SMC for miniaturization at higherfrequencies are done.In this paper, a compact lowpass filter having a 3-dBcutoff frequency of 10 GHz is designed. The employeddesign is the coplanar waveguide technology (CPW) forits effectiveness at high frequencies.The organization of the paper is as follows: First, inSection II, the design of a hybrid lowpass filter having acutoff frequency around 10 GHz is presented. Fabricationand measurement results are presented and discussed inSection III. Then, a comparison of the designed filter withcommercially available lowpass filter is made.Abstract—The paper presents the design of a compactcoplanar waveguide lowpass filter in the X band. Thelowpass filter has a 3-dB cutoff frequency of 10 GHz. Thecompact size is achieved due to the use of localized surfacemount capacitive loading. For the first time, theemployment of localized loading capacitors for miniaturization proves to be efficient at high frequencies. The designedhybrid filter proves to have smaller size and betterperformance than commercially available filter with whicha comparison is made. Index Terms—Coplanar waveguide technology, compactlowpass filter, miniaturization, semi-lumped, X-BandI. INTRODUCTIONIn microwave communication systems lowpass filter isan important device that is usually employed in thedesign of RF systems. Planar filters are very popularbecause they can be fabricated using PCB technology andare appropriate for commercial applications due to theirrelatively simply design, compactness and low cost offabrication.With the advancement of applications of wirelesscommunication systems, the development of filters hasemphasized compact size and high performance. Severalminiaturization techniques have been reported. The mostused techniques can be classified under three categories:the use of high permittivity substrates [1]-[5], tilting,folding and meandering [6]-[7], and the use of hybridapproach [8]-[15]. The three categories are discussed inthe next paragraphs.The use of substrates with high relative dielectricpermittivity ( r 20) is effectively reported in someresearches providing a good percentage of miniaturization of around 45% [1]-[5]. However, the drawbacks ofusing high permittivity substrates are bandwidthreduction, high cost, and difficulty in achieving highimpedances. High impedances are very important inachieving acceptable electrical characteristics.Meandering technique, on the other hand, helpsobtaining compact devices. However, the disadvantage ofsuch a technique is the increase of circuit complexity [6][7]. Moreover, parasitic coupling effects of nearbyelements should be accounted for while designing whichII. HYBRID LOWPASS FILTER DESIGNA. Employed TopologySeveral topologies based on the use of semi-elementapproach have been proposed. One of the efficienttopologies in reducing the size of a filter is the oneproposed in [8]. The authors in [8] proposed to loadManuscript received July 10, 2019; revised September 12, 2019;accepted October 1, 2019.Corresponding author: Hamza O. Issa (email: h.issa@bau.edu.lb). 2020 Int. J. Elec. & Elecn. Eng. & Telcomm.doi: 10.18178/ijeetc.9.2.105-110105
International Journal of Electrical and Electronic Engineering & Telecommunications Vol. 9, No. 2, March 2020transmission line sections at their middle by SMCcapacitors. The equation giving the value of the loadingcapacitance in function of the transmission lineparameters is available in [8]. Several lowpass filterswere designed at 1 GHz, fabricated and validated. Thefilters show good characteristics in terms of compactness,return loss, insertion loss, shape factor (selectivity) andsuppression of spurious. This topology is employed inthis paper to design a lowpass filter having a 3-dB cutofffrequency of 10 GHz. However, at high frequencies SMCcapacitors present parasitic effects. In the next subsection, a model of these capacitors is suggested andvalidated in order to use during filter design.constructed using the circuit simulator advanced designsystem (ADS). When simulated for its frequencyresponse, the capacitor circuit model will exhibit aminimum impedance value at a specific frequency calledthe self-resonant frequency (SRF). The SRF can beanalytically determined from the capacitance C and theseries parasitic inductance (Ls). At the SRF, the capacitiveand inductive reactance values are equal (1/2πfC 2πfLs).Capacitors behave as inductive devices at frequenciesabove the SRF and, as a result, cannot be employed. Themodel of capacitors is extracted using measurements byfitting the values of the parasitic elements Rs and Ls ofFig. 1.For measurements, the SMC capacitor is welded at thecenter of a CPW transmission line between the mainconductor and the ground plane as shown in Fig. 2. Theused SMC capacitor has a capacitance of 0.2 pF and acase size 0603. This capacitance value is of interest forthe design of the LPF in the next subsection. The usedCPW transmission line has a moderate characteristicimpedance of 50 . These arrangements guaranty goodaccuracy in extracting the model. It is worth recalling thatwithout accurate modeling, the use of the SMC capacitorswill induce design imprecision.B. Modeling of SMC CapacitorsAt high frequencies, the parasitic effects of lumpedelements (SMC capacitors here) must be taken intoaccount. Above a certain nominal frequency to be definedlater, lumped elements could introduce severe designproblems. Moreover, the pads used for soldering thecapacitors should be modeled for the design to beaccurate.Several models of SMC capacitors were reported byresearchers [16]-[18]. In this work, the series RLC modelis used. The model is inspired from different models in[16]-[18]. It is simple and efficient, as shown in Fig. 1.For each SMC capacitor, a parasitic series resistance Rsand an inductor Ls together with a shunt resistance (Rp)are used to create the model. These elements take intoaccount the parasitic effects at high frequencies. Theseries inductor Ls is included to represent the conductionand displacement current densities in the metallic anddielectric parts of the capacitor that lead to a surroundingmagnetic field. Rs is added to represent the dielectric andOhmic losses in the capacitor and is usually referred to asthe equivalent series resistance (ESR). The shuntresistance (Rp) represents the coupling losses between theelectrodes of the capacitor. Rp has usually very largevalues even at high frequencies. For simplifying themodel, Rp can be neglected. The measurement results willconfirm the validity of the last decision.Fig. 2. Photograph of the soldered capacitor.A TRL (thru, reflect, line) calibration was performedprior to measurements. The S-parameters of the loadedtransmission line were extracted up to 13.6 GHz using theRohde&Schwarz ZVL vector network analyzer.The next step was to fit the measurement results to thesimulation results of the model by varying the values ofRs and Ls. Fig. 3 shows a comparison of the S-parametersobtained from measurements and from model simulation.The best fit can be obtained for Ls 0.78 nH and Rs 530m . It is worth noticing that the SRF of these capacitorsdoes not appear in the investigated frequency band. So,this type of capacitors can be used safely for targeted Xband frequency range. The same procedure is done foranother capacitance value.This value is 0.15 pF. The results of the capacitor’smodel optimization are shown in Fig. 4. The figure showsa comparison of the S parameters results betweenmeasurements and model simulations of the 0.15 pF SMCcapacitor (case size 0603).Fig. 1. Capacitor equivalent model.As for the pads used for soldering the SMC capacitors,we propose to model them as short transmission lines.The length of the transmission lines representing the padsis kept less than /10. This choice is very realistic sincethe pads are extensions of lossy transmission lines inmost cases. Neglecting the parasitic effects of the padsmodifies the propagative characteristics of the EM waveand hence the design.Fig. 1 shows the complete used parasitic model of theSMC capacitor having a nominal capacitance value C(pF). In this case, the capacitor is connected in seriesbetween an input and an output port. The model is 2020 Int. J. Elec. & Elecn. Eng. & Telcomm.106
International Journal of Electrical and Electronic Engineering & Telecommunications Vol. 9, No. 2, March 20200-10-20Modeled S21(dB)Measured S21(dB)-30Modeled S11(dB)for Ls 0.78nH-40Measured S11(dB)-5050105151015Freq(GHz)Freq(GHz)Fig. 3. Comparison of S-parameters of the 0.2 pF capacitor: Measuredand modeled.0-10-20Modeled S21(dB)-30Modeled S11(dB)for Ls 0.78nHMeasured S21(dB)Measured S11(dB)-40-50510Freq(GHz)01551015Freq(GHz)Fig. 4. Comparison of S-parameters of the 0.15 pF capacitor: Measuredand modeled.Again, this capacitance value is of interest for thedesign of the LPF in the next subsection. The obtainedvalue of the series inductance Ls was found to be 0.78 nHand the value of the equivalent series resistance Rs is 580m . Ls has the same value as in the previous case of the0.2 pF capacitor. This is logical because theoretically thevalue of Ls is defined by the case size and dimensions ofthe employed SMC (0603 in this case). Here again, theSRF does not appear in the band of interest. On the otherhand, Rs which represents the dielectric and Ohmic lossesin the capacitor (usually referred to as the equivalentseries resistance ESR) depends on the value of thecapacitance and defined by the fabrication accuracy ofthe capacitor.C. Lowpass Filter DesignThe desired filter template is of lowpass nature havinga 3-dB cutoff frequency at around 10 GHz. The designedfilter will be compared to commercially available filterRLPF13G09 offered by offered by RF-LAMBDA .For comparison reasons, the desired lowpass filtertemplate should exhibit an insertion loss of no more than1 dB at half the cutoff frequency fc, and the spuriousshould not appear until at least double the cutofffrequency. The filter should be matched (return lossbetter than -10 dB within the operating passband) andhave a cutoff roll selectivity higher that 50 dB/GHz. Thedesign encompasses both distributed elements and 2020 Int. J. Elec. & Elecn. Eng. & Telcomm.localized ones. As the target is to operate in the X-band,the localized component is chosen to be capacitor. Thereason is that capacitors have relatively higher qualityfactors than inductors. The idea, again, is to transfer theModeleduse S21(dB)of lumped SMC elements into higher frequency usingappropriateMeasuredS21(dB) design (technology and topology).Thecoplanar waveguide (CPW) technology is adopted.Modeled S11(dB)reasonfor using the CPW technology is because itfor LThe 0.78nHshas S11(dB)generally smaller frequency dispersion thanMeasuredmicrostrip technology especially in the case of our designwhere narrow transmission lines (high characteristicimpedances) are required. Moreover, the presence of theground planes from both sides of the strip in the case ofCPW transmission lines avoids the use of via holes that,at relatively high operating frequencies, need to beprecisely modeled. The filter is designed using RogersRO4003 substrate having as parameters: dielectricconstant εr 3.55, height h 813 µm, metal thickness t 35µm and loss tangent tan 0.0027.Fig. 5 shows an ideal schematic of the proposed filter.It is composedof six sections defining, hence, the orderModeledS21(dB)of the S21(dB)filter (sixth order). This order is sufficient to get aMeasuredhigh roll off selectivity as required. Each elementaryModeled S11(dB)is composed of two ideal transmission lines and aforsectionLs 0.78nHcapacitor.The capacitor is connected at the center of theMeasured S11(dB)transmission lines for simplicity reasons. The parasiticmodel of the capacitors is not considered is thispreliminary investigation.In order to get good matching characteristics in thepassband, taperization is used as shown in Fig. 5.Taperization consists of breaking the topology’speriodicity by changing the characteristic of the sectionsat the input and at the output of the filter. Taperization isachieved in this case by varying the length of thetransmission lines at the near and far end sections ( 1, L1)and/or varying the corresponding loading capacitance ofthe near and far-end sections (C1). On the other hand, thesections at the middle have equal electrical length 2(corresponding to physical length L2) and loaded bycapacitors of same capacitance value C2. It is worthmentioning that the characteristic impedance Zc of alltransmission line sections (middle and near and far ends)have the same value. For minimal filter dimensions, it isinteresting to use high characteristic impedance values.However, due to fabrication constrains the maximumcharacteristic impedance that can be used for all thetransmission lines is limited in this work to 120 Ohmscorresponding on the considered substrate to a width (W)of 0.25 mm and a gap (G) of 1.3 mm. For ground planeuniformity, all transmission lines will have the samevalues for the width W and the gap G.Fig. 5. Schematic of the sixth order lowpass filter.107
International Journal of Electrical and Electronic Engineering & Telecommunications Vol. 9, No. 2, March 20200-10-20-30-40simulation results are shown in Fig. 8. The cut-offfrequency is at 9.7 GHz, the return loss is -16 dB, thespurious appears at 25 GHz, and the insertion loss is 0.11dB at half the cut-off frequency and remains less than 0.5dB until about 9 GHz. Moreover, the filter’s cutoff rollselectivity is 65.8 dB/GHz. As the desired template is met,the filter is fabricated and measured for validation.S21 (dB)0-50S21 (dB)S11 07080Freq (Ghz)S11 (dB)-70-80-90-10001020304050607080Freq (Ghz)Fig. 6. Preliminary results of the return loss S11 and the insertion lossS21 in dB of the lowpass filter.12.88Fig. 8. Momentum Simulation Results.Ground PlaneC1/20.77510.25Pad1.50.25L2 L10.8Ground Plane(a)Fig. 7. Layout of the CPW lowpass filter (dimensions in mm).(b)Fig. 6 shows the result of the first optimization thatdoes not take into account all fabrication constrains anddifferent parasitic effects. The optimized values of L1, L2,G, W, C1 and C2 are 0.96 mm, 1.29 mm, 5 mm, 1.64 mm,0.42 pF and 0.49 pF respectively.The simulation results meet the desired template. Thematching in the passband is better than 20 dB and thereare no spurious in the large band response. However,these results are very optimistic since there are someimportant effects that should be taken into account. Asdiscussed earlier, at high frequencies, the parasitic effectsof capacitors and the pads used for soldering them mustbe taken into account. Moreover, the capacitance valuesmust be normalized.The filter is re-optimized to meet the desired templateby modifying the lengths of the transmission lines and thevalues of C1 and C2. The new optimization includes thecapacitors’ and pads’ models and all fabricationconstraints. The optimization is done using theelectromagnetic simulator Momentumof ADS as it yields-5more accurate results. The generated layout of theoptimized lowpass filter is shown-15in Fig. 7 (dimensions inmm). For clarity, not all the capacitors are shown in the-25 represented. As shownfigure. Only C1 at the near end isin Fig. 7 for C1, all capacitors are divided into two-35connected in parallel across the maintransmission line.Doing this ensures a more balanced response and helps-45reducing the effect of the series parasitic elements of thecapacitor. The expected advantageis lower insertion loss-55of the lowpass filter. The capacitors used are the same asthe ones modeled in the previous-65 section C1/2 0.15 pF(taperized sections) C2/2 0.2 pF (middle sections). TheIII. FABRICATION AND MEASURMENTSFig. 9 shows the fabricated filter ((a) and (b)) and theTRL calibration kit necessary for measurements (c).Access lines were added at the two ports of the filter formeasurement purposes. Fig. 9 (a) shows the filter withoutpackaging. Fig. 9 (b) shows the measured filter withpackaging.Measurements were done using the ROHDE &SCHWARZ vector network analyzer up till 13.6 GHz.Fig. 10 shows that there is a good agreement betweenmeasurements and SimulatedS11(dB)-750022-7500 2020 Int. J. Elec. & Elecn. Eng. & Telcomm.224466446688101012121414Fig. 10. Measured and simulation results of the return loss S11 and theinsertion loss S21 in dB.-65-75(c)Fig. 9. Fabricated devices: (a). the realized hybrid lowpass filter(with access lines ), (b). the tested filter with packaging, (c).TRL calibration set.88108101012121414
International Journal of Electrical and Electronic Engineering & Telecommunications Vol. 9, No. 2, March 2020The good electrical performance of the filter and itscompactness are due to the use of SMC elements. Theseelements have to be well modeled in order to beemployed at high frequency.The results in Fig. 10 shows that SMC loadingapproach has potential in designing compact filters athigh frequency (around 10 GHz) without deteriorating theelectrical performance of the device. For that, surfacemounted capacitors can be used in conjunction withcoplanar waveguide technology offering, thus, very goodcompactness, maintained performance, and reduced cost.For performance evaluation, the designed filter iscompared with a commercially available one. The chosenfilter is available among the products offered by RFLambda. The filter chosen is the RLPF13G09. Thetechnology used in designing the filter is the SuspendedStripline technology. Of course, this technology is moreexpensive and relatively complicated. The filter iscompared, as well, with other state-of-art filters [19]-[21].IV. CONCLUSIONThe design of a compact coplanar waveguide lowpassfilter in the X band was presented. The lowpass filter hasa 3-dB cutoff frequency of 10 GHz. The compact designmakes use of localized surfac
coplanar waveguide lowpass filter in the X band. The lowpass filter has a 3-dB cutoff frequency of 10 GHz. The compact size is achieved due to the use of localized surface mount capacitive loading. For the first time, the employment of localized loading capacitors for miniaturiza-tion proves to be efficient at high frequencies.
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